ANALOG DEVICES Low Noise-Wideband Chopper Stabilized Amplifier FEATURES Low Drift: 0.1uV/C, 1IpA/C (Model 234L) Offset Stability: 2uV per month Submicrovolt Noise: 0.7uV p-p (0.01 to 1Hz B.W.} Fast Response: 2.5MHz B.W., 4us settling (0.01%) Low Cost Module Small Size: 1% x 1% x 0.4" APPLICATIONS Precision Wideband Amplification Current and Voltage Summation High Speed Integration Reference Buffering Controlled Current Source Bridge Amplifier GENERAL DESCRIPTION Analog Devices model 234 is a high performance chopper sta- bilized op amp which significantly improves on the noise and bandwidth performance of previous designs. Available with drift of 0.1uV/C, the model 234 features 0.7uV p-p input noise and 2.5MHz unity gain bandwidth to satisfy many de- manding requirements for a premium amplifier at less than premium prices. Incorporating MOSFET choppers and discrete components (vs. IC op amps) for the main and stabilizing amplifier chan- nels, this inverting design is virtually free of input chopper spikes and offers reduced modulation ripple for quieter wide- band performance. These characteristics are especially desira- ble when operating from high source impedances (above 100kQ2) at wide bandwidths. To illustrate the improvements in noise and bandwidth performance, over previous Analog De- vices designs, comparative data is set forth in the following sections comparing models 232 and 233 with 234. Other model 234 specifications include: gains of 107 V/V, 4us settling time to 0.01% (20kQ load, 10V) and three selections for voltage drift: vec (234J), 0.3uV/C (234K), and 0.1pV/C (234L). Available in a compact plug-in module (1%" x 14%" x 0.4"), model 234 is competitively priced for new OEM designs and is recommended as a pin compatible re- placement for upgrading the performance of most existing de- signs. The use of premium discrete components throughout assures repeatable unit-to-unit performance for best results at lower costs. APPLICATIONS In general, the model 234 inverting amplifier should be con- sidered where long term stability of offset voltage must be Ey ANALOG DY maintained with time and temperature for precision designs, or wherever carefree operation of instruments and remote cir- cuits is essential. Typical applications include low drift amplifi- cation of wideband microvolt signals, integration of low duty- cycle pulse trains and fast analog computing for general pur- pose designs. Low input noise and stable offset voltages also make model 234 an ideal preamp for precision low frequency applications such as DVMs, 12 to 16 bit A to D converters, and for error amplifiers in servo and null detector systems. IMPROVED NOISE AND BANDWIDTH PERFORMANCE The improved performance of model 234 accrues from the use of discrete components throughout, coupled with low noise front-end circuits, all carefully packaged and shielded to mini- mize pickup and intermodulation effects. Chopper modulation ripple, as shown in Figure 1, is significantly reduced over an earlier design, model 232, for most wideband applications. 250k 10k L 234 Figure 1. Comparative Input Noise (RTI} Performance in adc to 1kHz Bandwidth OPERATIONAL AMPLIFIERS 81SPECIFICATIONS (typical @ +25C and +15V unless otherwise noted) MODEL 234) 234K 2341, OPEN LOOP GAIN DC , 2k ohm load 107 V/V min * * RATED OUTPUT Voltage +10V min * * Current +5mA min * * Load Capacitance Range 0-1000pF min * * FREQUENCY Unity Gain, Small Signal 2.5MHz * * Full Power Response 500kHz min * * Slew Rate 30V/ps * * SETTLING TIME to 0.01% 20kQ load, 10V step (Figure 2) 4us * * INPUT OFFSET VOLTAGE Initial Offset! @ +25C 50uV max +20uV max +20uV max vs. Temp, 0 to +70C +1.0uv/C max +0.3uV/C max +0.14V/C max vs. Supply Voltage 0.2UV/% * * vs. Time +2uV/roonth * * vs. Turn On, 10 sec to 10 min +3yuV * * INPUT BIAS CURRENT Initial, @ +25C vs. Temp, 0 to +70C vs. Supply Voltage +100pA max +4pA/C max +0.5pA/% co * t2pA/C max t1pA/C max * * INPUT IMPEDANCE Inverting Input to Signal Ground 300k ohms * * INPUT NOISE Voltage, 0.01 to 1Hz 0.7UV p-p * * 0.1 to 10Hz 1.5uV p-p * * 10Hz to 10kHz 2nV rms * * Current, 0.01 to 1Hz 2pA p-p * * 0.1 to 10Hz 4pA p-p * * INPUT VOLTAGE RANGE {-) Input to Signal Ground +15V max * * POWER SUPPLY (V dc)? Rated Performance Operating +15V @ SmA +(12 to 18)V TEMPERATURE RANGE Rated Specifications Operating Storage 0to 470C 5 -25C to +85C -25C to +100C *Specifications same as model 234). Externally adjustable to zero. ? Recommended power supply, Analog Devices model 904, +15V @ 50mA Specifications subject to change without notice. 82 OPERATIONAL AMPLIFIERS OUTLINE DIMENSIONS Dimensions shown in inches and (mm). j> 151 MAX (38.1) +| oat max? 10.4) a __U 0.04 DIA (1.02) =| | 0.20 MIN (5.0) 0.25 MAX (6.4) 1.51 MAX (38.1) 0.1 GRID (2.54) BOTTOM VIEW NOTES: *Optional Trim Pot Analog Devices Model 79PR50k Connect Trim Terminal to Common if Trim Pot is not used. 1. SG (SIGNAL GROUND) Tied to Common. 2. Mating Socket AC1010 3. Weight: 27 grm OPEN LOOP GAIN AND PHASE SHIFT PHASE SHIFT OPEN LOOP PHASE SHIFT DEGREES 10 10 10 FREQUENCY -- Hz 10kQ vw JT 10V 0 ~ - OUT >] one IV a = 20k{2 Figure 2. Settling Time Test Cir- Cuit Using Scope Comparator Preamp at AShown below are plots of typical input voltage and input current noise over the frequency range of 0.01Hz to 10Hz. Particular care has been exercised in the design of this amplifier to reduce the noise level to that commensurate with the low drift performance obtained by chopper stabilization. En kV { 0.01 - 10Hz 10 SEC he i | | y 11 | 0.2pA{ | In TT ay 0.01 - 10Hz Figure 3. Model 234 Voltage and Current Noise INPUT IMPEDANCE CONSIDERATIONS The maximum input impedance for inverting amplifiers of all types is limited by bias current, bias current drift, and noise current. These currents flowing through the source impedance will increase the total error and noise when the input imped- ance exceeds E/I, where E is a given type of voltage error and I is the corresponding current error. Figure 4 is a plot of total offset voltage, total voltage drift and total noise vs. input re- sistance for the model 234. Up to 100,000 ohms, the model 234 provides relatively constant levels of offset, noise, and drift. Above this resistance level, the bias current effects be- come more predominant. 100 VOLTAGE FT (uv/C) 234) OFFSET (uV} 0.01 - 10Hz (uV p-p) ALL MODELS NOISE 234K/L OFFSET (uV) 234L VOLTAGE IFT (xv, ALL MODELS, {2.1 - THz INPUT VOLTAGE: NOISE, DRIFT, OFFSET (aV p-p or uV/C) 234K VOLTAGE 234) AGE ORIFT (uV/C)} o.1 104 108 106 107 Rin onms Figure 4. Uncompensated Offset, Drift and Noise vs. Rijn INITIAL OFFSET ADJUSTMENT A valuable characteristic of the model 234 is the low offset voltage without external trim. The specification is 5}04V maxi- mum for the model 234J, and 20uV maximum for the models 234K and 234L. In many applications there will be no need to zero the offset since it is so low. In such cases the trim ter- minal may either be left open, or grounded, whichever is more convenient for the user. If voltage offset adjustment is desired, it may be done with a potentiometer or selected fixed resistor network, as shown in the outline drawing on previous page. Input bias current flowing through the input resistor(s) creates additional voltage offset, particularly with input resistances ex- ceeding 500,000 ohms. For circuits where the total input and source resistance remain relatively constant, the entire offset may be zeroed out with the voltage offset adjustment. No additional drift will occur with the model 234 when voltage trimming is used to compensate for the offset effects of input bias current. The circuit of Figure 5 should be used to compensate for bias current offsets when using the model 234 as a current to volt- voltage converter. The potentiometer-resistor network pro- vides a compensating bias current to cancel the amplifiers own input bias current. The offset voltage trim may be used but is not necessary when using this technique. When the amplifier is used with a widely varying input resist- ance and minimum offset is desired, the voltage and current trim potentiometers should be used. The voltage offset should be zeroed with a low value (e.g. 1k ohm) resistor connected from the inverting input to ground. The offset current adjust- ment should be made with the maximum expected value of Rj connected between the input and ground. lin Re bo Eout = (R/Ri) Ejn VOLTAGE OFFSET ADJUST (50kS2 pot) CURRENT OFFSET ADJUST (50kQ. pot) v Figure 5. Offset Current Voltage Cancellation 10M22 INVERTING OPERATION The model 234 is designed for use in the inverting mode. it is important that the SG (equivalent to +in) terminal be kept at the same potential as the amplifiers common terminal. Any voltage difference between these points is similar to a common mode voltage, and performance cannot be guaranteed under such conditions. The model 234 is also an excellent amplifier for measurement and conversion of low level current sources to proportionate voltages. With offset current externally zeroed, input currents of ten to twenty picoamperes can be amplified and converted to a voltage source for further processing. SHIELDING, PICKUP AND GROUNDS A special feature of the model 234 is the internal electrostatic static shield. This prevents not only pickup of extraneous signals by the module but also prevents radiation of chopper noise by the module. One precaution is to insure that noise sources are shielded from the inverting input. The user should OPERATIONAL AMPLIFIERS 83also insure that ground loops do not occur which can add ex- traneous signals when amplifying from microvolt or millivolt sources. Figure 6 illustrates the proper connections to avoid ground loops. AW Ry R, NAA PN SG = SIGNAL qd a RL RETURN L LEAD x gtOAD RETURN POWER SUPPLY COMMON * SIGNAL RETURN AND LOAD RETURN SHOULD BE CONNECTED FO POWER COMMON AS CLOSE TO AMPLIFIER PINS AS POSSIBLE Figure 6. Ground Connection INTERMODULATION CONSIDERATIONS If noise at medium frequencies (to 400Hz) finds its way into the input circuits of carrier amplifiers (chopper amplifiers and the chopper-stabilizing portions of chopper-stz.bilized ampli- fiers), it tends to beat with the chopper frequency and pro- duce sum and difference frequencies. The sum frequencies are unimportant, because they are usually filtered out; the noise frequency is usually unimportant because it, too, is filtered out. But the difference frequencies (which can include dc) usually interfere directly with the low-level low-frequency signal information. There are precautions that can be taken by the manufacturer to minimize such interference occurring within the devices themselves; but the user must also be aware of the need for precautions, especially in performing low-level measurements in the presence of: 1. input signals containing high-frequency :ormal-mode noise components (such as unfiltered ca:rier from a measuring device) 2. ripple coupled in from power supplies 3. stray electromagnetic radiation at line frequencies, especially if it is rich in harmonics. This noise may be introduced to the amplifier at either impro- perly guarded input leads or at the power supply terminals. These effects may be minimized by using shielded supplies which have low ripple and low source impedances at the line harmonics. Properly shielding the input leads, as well as locating the amplifier as far from sources of 60Hz (or 50Hz) magnetic fields, is also recommended for best performance. Mechanical orientation of the amplifier package and layout of signal grounds may also be used to minimize EMI effects. If a beat does occur, it usually manifests itself as a slowly varying offset signal at the output of the amplifier, usually below 20Hz. To examine the extent of this equivalent offset noise voltage in a system, an oscilloscope should be used to monitor the amplifier output with the input signal point shorted to ground. As another test, low leve! signal may be applied at the input of the final circuit configuration to deter- mine the intermodulation rejection capability of the design. In this test, the signal frequency should be swept through the modulation frequency point to observe output signal peaking. A low pass output filter, at approximately 40.Hz, should be used when making these tests. THE T NETWORK High gains and high input impedance to an inverting amplifier normally require excessively large feedback resistors. For ex- 84 OPERATIONAL AMPLIFIERS ample, an input impedance of 1,000,000 ohmis and a gain of 100 require a feedback resistor of 100 Megohms. Such a re- sistor is relatively expensive, particularly for low tolerance units. Furthermore, one picofarad of stray capacitance across this single resistor would reduce 3dB bandwidth to 1590Hz, and resist.ve leakage across PC boards may become a problem. The T network in Figure 7 is a means of minimizing these problems. If the ratio R/R; is at least 5 to 1, there will be no measurab.e change in other performance characteristics. If the ratio is lower, for instance, 1 to 1, the effective drift and noise gain will be doubled, compared to the signal gain. A general rule is to make the ratio R/Rj approximately equal to the ratio R,/R,. This normally results in reasonable values of resistance for R, and a minimal increase in noise and drift gains corm.pared to the standard two resistor circuit. An addi- tional advantage of the T network is variable gain without the necessity of connecting a switch or potentiometer directly to the highly sensitive inverting input terminal. This avoids serious noise pickup problems. In such a hookup, R, is the variable element. Ry Rg : Re Ry Rai] [R2eiRa Re ; (SG) oR Ry Ht Ry " com Figure 7. T Network OVERLOAD RECOVERY The overload recovery circuit shown in Figure 8 will prevent the input circuitry from becoming saturated. This circuit, con- nected externally, will allow the amplifier to recover from overload in less than 0.5us. Without this circuit overload re- covery will require up to 5 seconds. [rast RECOVERY 15k J cracuit pM. fe | Ss T a INDO 1B >. USE 1F OVEFILOAO CURRENTS IN EXCESS OF AMPLIFIER . OUTPUT CURRENT CAPABILITY MOY BE ANTICIPATED fl Figure 8. Overload Recovery Circuit HIGH SOURCE IMPEDANCE CIRCUITS When required to operate from source impedances above 100k9, the model 234, with inherently lower input current noise and spikes, offers dramatic improvements over previous designs. (See Figure 9) 300k!) 3Mf? 10k!? a loHz 7 Filter C _ Figure 9. Comparative Input Noise (RTI) Performance ina de to 10Hz Bandwidth MODEL 232 _ MODEL 233 ~ ' + suv ATi | af 5 SEC MODEL 234 L J] 1